The present invention relates to a dual branch receiver.
Dual branch receivers well known. One example is described in U.S. Pat. No. 4633315. The interest in dual branch receivers stems primarily from the desire to manufacture as much as possible of a radio receiver as an integrated circuit. A problem in making a receiver circuit by integration is the construction of the filter circuits because it has been long recognised that it is not easy to build a band-pass filter whose pass-band is very narrow. N. F. Barber in an article "Narrow Band-Pass Filter Using Modulation" Wireless Engineer, May 1947 pages 132 to 134 discloses a filter in which an incoming signal is supplied to two branches each including three stages constituted by (1) a modulator for frequency down-converting the incoming signal, (2) a low pass filter for passing the difference component of the modulation and (3) another modulator for frequency up-converting the low pass filtered signal. The modulators in the two branches of the first and third stages are quadrature related with respect to each other. The outputs of the modulators in the third stages are recombined to provide a narrow band-pass filtered signal. Barber discusses the effects of errors resulting from the phase splitting and the differences in gains in the two branches. In the case of using this type of filter in an FM receiver in which after the first stage the modulation is folded about zero frequency, mismatching of gain and deviation from orthogonality between the two signal branches can give rise to an unwanted image being generated. This will result in distortion, and possibly a whistling tone, in the demodulated audio output.
FIG. 1 of U.S. Pat. No. 4633315 shows a dual branch receiver for television signals, comprising first and second branches I and Q, respectively, constituted by first and third mixers and second and fourth mixers. A signal input terminal is coupled to the first and second mixers, each of which also receives a respective one of the in-phase and quadrature phase outputs of an r.f. local oscillator. The r.f. oscillator frequency (fl.sub.1) is offset by fo from the input carrier frequency (fc), fo having a value of the order of 100 to 200 Hz. The input signal is mixed down to a baseband, is low pass filtered and in the third and fourth mixers is frequency up-converted using the quadrature related outputs of a second intermediate frequency local oscillator. The in-phase and quadrature-phase signals in the first and second branches are applied to a scanning circuit and a differencing circuit by which the video and sound signals can be obtained.
In U.S. Pat. No. 4633315 gain and phase control are provided to correct for the imbalances between the two paths. Error signals for use in the control are derived from deviations in amplitude and phase of the unwanted image components at the outputs of the summing and differencing circuits. The particular embodiment disclosed in FIG. 1 of the patent uses the picture (or video) carrier signal V as a reference. This carrier signal is applied to a narrowband phase locked loop (PLL) which produces as detected carriers an in-phase carrier and a 90.degree. out-of-phase carrier. These two detection carriers are applied to respective synchronous demodulators which also receive the output V of the difference circuit. The outputs of the synchronous demodulators are low pass filtered to provide d.c. voltages. The d.c. voltage derived using the in-phase carrier from the PLL is applied to an amplitude control circuit which controls the mixing gain of the fourth mixer by amplifying the oscillator mixing signal applied thereto. The d.c. voltage derived using the 90.degree. out-of-phase PLL signal is used to effect phase control by varying the phase quadrature relationship between the two second (I.F.) oscillator mixing signals.
Introducing the small offset frequency fo between the first, r.f., local oscillator frequency fl.sub.1 and the input carrier frequency fc enables separation of the unwanted image signal Du to be separated from the desired signal Dw. FIG. 1 of the accompanying drawings shows the signal components at the outputs of the differencing circuit, that is I-Q, assuming that the transmitted carrier fc is unmodulated. In this drawing, the wanted signal component Dw is located at the frequency fl.sub.2 +fo, where fl.sub.2 is the frequency of the second local oscillator, while the unwanted image component Du is at fl.sub.2 -fo. The wanted and unwanted components can be easily distinguished.
The situation at the output of the differencing circuit becomes more complex if the transmitted carrier is frequency modulated. When modulating the signal there are more signal components at the harmonics of the modulating frequency, and their levels are governed by the modulation index employed. A problem arises however if the modulation frequency fm and the offset frequency fo become equal because the harmonics of fm in the wanted signal component Dw will fall on top of the image component Du. As a result it is impossible to detect the image component Du independently of the wanted component Dw in the output of the differencing circuit. If this occurs, the compensation schemes for the deviations in gain and phase between the I and Q branches, for example as described in U.S. patent specification 4633315, will cease to function correctly because they depend on the unwanted image component Du to provide the necessary error signal. Consequently the demodulated output from the receiver will suffer from distortions.